Tuning control arrangement

ABSTRACT

A tuning arrangement for a resonant circuit, the tuning arrangement having an output reactance dependent upon a plurality of input applied signals one of which is an input tuning signal, and including: an array of tuning circuits connected in a network whose output reactance is used to control the resonance frequency of the resonant circuit, each tuning circuit having a control input and having a reactance which varies in dependence upon the value of a control signal applied to the control input; elements for generating a plurality of different control signals for application to the control inputs of the tuning circuits; and each such control signal varying substantially linearly with the input tuning signal throughout a predetermined range specific to that control signal, so that the frequency response of the resonant circuit to the input tuning signal is substantially linear throughout a desired range of the input tuning signal.

FIELD OF THE INVENTION

The present invention relates to an arrangement and a method forgenerating an input signal for a tunable circuit such as an oscillator,resonator or filter circuit.

BACKGROUND OF THE INVENTION

Many frequency controlled tunable circuits, such as oscillator or filtercircuits, have non-linear frequency output characteristics, that is, thefrequency output vs. control (current or potential) characteristic isnot always linear. When good linearity is desired, oscillator designerstypically select component types and apply resonant effects to achieve acharacteristic that meets the requirement as nearly as possible.Sometimes it is not possible or practical to achieve the desiredlinearity and system designs need to be modified. Even when linearity isachievable, component and dimension tolerances can require either thatcomponents are individually selected following device tests, or thatphysical adjustments are made during the manufacturing process. Thisobviously increases production costs.

In some applications, absolute control rate can be important.Temperature dependencies can limit the final system performance, orrequire that an otherwise low-power system be placed in a temperaturecontrolled environment, such as an oven.

A constant frequency tuning rate can be important for systems such astemperature compensated crystal oscillators, to avoid degradation offrequency accuracy when the oscillator is tuned away from the conditionsunder which it was compensated. Such tuning may be necessary to correctfor, for example, ageing of the quartz crystal, or to match theoperating environment. One situation where this is particularly relevantis in retiming circuits where a local crystal oscillator tracks anintermittent input clock with a frequency that is allowed to deviatefrom a nominal value. To ensure a rapid recovery from loss of inputclock signal, the local crystal oscillator is required to continueoscillating at the last observed frequency for extended periods.

Communication, navigation, and timing systems often need to synchronisetheir local frequency sources to a remote reference. It is helpful forthe control of this synchronisation for the tuning rate of the localsource to be constant—i.e. linear and independent of temperature. Thesesame characteristics are beneficial for maintaining constant frequencyoutput from temperature compensated oscillators after the oscillator hasbeen re-tuned. Suitable resonators for such oscillators can includeacoustic devices such as bulk-mode crystals and SAWs, dielectricresonators such as ceramic pucks and cooled sapphire and hybridarrangements.

There is thus a need to provide for a more linear tuning rate for suchsystems, and/or for the tuning rate to be matched for individualoscillators or resonators and/or to provide compensation for typicaltemperature variations in oscillator output.

In GB 2369259 some tuning linearity and temperature corrections can beachieved using a circuit which provides a predistorted control signal.While the method of predistortion in GB 2369259 was capable of providingtuning characteristics with excellent linearity and temperatureindependence, it can prove difficult to maintain fast response and lownoise while minimizing the circuit's dissipation. The present inventioncan provide an alternative and improved way of achieving linearity andtemperature correction.

BRIEF SUMMARY OF THE INVENTION

According to one aspect of the invention there is provided

a tuning arrangement for a resonant circuit, the tuning arrangementhaving an output reactance dependent upon a plurality of input appliedsignals one of which is an input tuning signal, and comprising:

an array of tuning circuits connected in a network whose outputreactance is used to control the resonance frequency of the resonantcircuit, each tuning circuit having a control input and having areactance which varies in dependence upon the value of a control signalapplied to the control input;

means for generating a plurality of different control signals forapplication to the control inputs of the tuning circuits; and

each such control signal varying substantially linearly with the inputtuning signal throughout a predetermined range specific to that controlsignal,

so that the frequency response of the resonant circuit to the inputtuning signal is substantially linear throughout a desired range of theinput tuning signal.

A corresponding method of tuning a resonant circuit is provided in whichan input applied signal is split to generate a plurality of individualcontrol signals of different values, each different individual controlsignal is applied to a control input of a tuning circuit in an array oftuning circuits, each of whose output reactance varies in dependenceupon the value of the control signal applied to its control input.

This method may use the tuning arrangement of the first aspect and canbe used for tuning a quartz crystal oscillator.

The network of tuning circuits may be a parallel network.

Preferably means are provided for applying a plurality of input signalsto adjust any second and third order nonlinearities in the tuning of thearrangement. The number of input signals required is less than thenumber of tuning circuits in the array. The input signals may beindependent of each other and may be modified using pre-set values forexample as a function of temperature to correct for temperaturedependence of a circuit to be tuned.

The means for generating the or each control signal may comprise apassive bias network, such as a potential divider arrangement comprisingan array of resistors.

In one embodiment the potential divider arrangement comprises a ladderformed by resistors arranged in two series chains of resistors formingthe sides i.e. the uprights of the ladder with a parallel array ofresistors interconnecting corresponding nodes in the two series chainsforming rungs of the ladder. Output would be taken from one side orupright of the ladder. Resistor values may be determined such thatcontrol potentials applied to the feet and the tips of the ladders serveto adjust the frequency offset, the sensitivity, and the second andthird order coefficients of the tuning law.

The values of the resistors in the sides and the rungs could be madesimilar to each other such that with N rungs in the ladder, the valuesof the individual rungs would be in the order N² larger than theindividual resistors in the sides.

Alternatively the divider arrangement may comprise two stars ofresistors with corresponding points joined to each other, and adjacentpoints additionally joined via a chain of resistors.

According to a preferred embodiment the tuning circuits comprisevariable reactance elements such as MOS varactors arranged in pairselectrically connected back-to-back at nodes with one terminal of thefirst element of each pair being electrically connected to theequivalent terminal of the second element of the same pair. Eachindividual control voltage is applied at the node between a respectiveone of the pairs of variable reactance elements.

Embodiments of the invention have the advantages of minimizingsensitivity to drift in the control characteristic of the variablereactance elements, linearizing tuning, maintaining tuning rate andlinearity with temperature, and allowing the linearity and the tuningrate to be adjusted variably over temperatures.

The invention may also serve to compensate for the effect of drift inthe control characteristics of individual tuning devices. Thearrangement can also minimize the sensitivity to variability in controlrequirements. This is because the individual tuning circuits are takenrelatively rapidly through the most sensitive parts of their tuningcharacteristics. In addition, in an arrangement of the invention whereall control signals have the same gradient, errors due to uniform shiftsin the responses (of the tuning circuits to the control signals) can becorrected by a signal- and temperature-invariant shift of the inputtuning signal. Such shifts are those of the control signal (required forany specific output) that are the same for all tuning circuits and areindependent of signal level and of temperature; such shifts areexperienced by MOSFET varactors under the action of radiation—or oflong-term aging, for example.

The tuning system of the present invention may thus be matched to theproperties of different resonators and oscillators, both by type andindividually.

The invention may also be used for maintaining the tuning rate of sometypes of R-C and ring oscillator arrangements, as well as of analoguefilters.

This invention may be used in combination with some methods of the priorart such as predistortion and the switched connection of differentdevices for different circuits to provide further improvements in thetuning rate, especially at the extreme of the available control range.

Thus the invention provides for controlling and adjusting the level ofoffset between tuning signals applied to individual tuning devices, andfor applying different gains between the input tuning signal and theindividual control signals applied to each tuning element.

In a general aspect the invention provides for applying several signalsthat are linearly dependent on an input signal as control signals to anarray of non-linear tuning circuits and the assembly is used to tune theresonance frequency of an electronic circuit. The characteristics of thetuning components (eg device areas if the tuning circuits are MOSvaractors), and the responses of the control signal to the input signalare arranged so that the resonance frequency of the circuit dependssubstantially linearly on the input signal.

DESCRIPTION OF DRAWINGS

For a better understanding of the present invention and to show how thesame may be carried into effect, reference will now be made, by way ofexample, to the accompanying drawings in which:

FIG. 1 is a graph showing the reactance characteristics versus appliedtuning voltage of a typical MOSFET varactor together with the reactancecharacteristics of a circuit arranged in accordance with an embodimentof present invention;

FIG. 2 is a schematic diagram of a circuit that can provide adjustmentof different orders of the tuning characteristic, arranged in accordancewith one embodiment of the present invention in which DC offset may beadjusted;

FIG. 3 is a schematic diagram of circuit comprising back-to-back MOSFETvaractors according to an embodiment of the present invention;

FIG. 4 a is a schematic diagram of a circuit for controlling theposition-dependent bias on a varactor chain according to an embodimentof the present invention;

FIG. 4 b is a graph showing the effect of modifying voltage sources inthe circuit shown in FIG. 4 a;

FIG. 5 is a schematic diagram of a circuit comprising an alternativeresistor topology for reference chains with adjustable distortionaccording to an embodiment of the present invention.

FIGS. 6 to 13 illustrate exemplary control signals applied to thearrangement.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1 shows how the present invention can be used to create a circuitwith improved reactance characteristics when compared to the impedance vbias of a typical MOSFET varactor.

In FIG. 1 curve 1 illustrates the non-linearity of the impedancecharacteristics of a typical MOSFET varactor. The horizontal axislabelled as time represents a linear voltage sweep. Curve 2 shows howthe present invention can provide significantly improved linearity inthe impedance characteristics. Curve 3 shows the difference betweencurve 2 and a linear response, showing that the impedance range overwhich this arrangement can have linear tuning exceeds a factor of two.Curve 2 shows the impedance of a parallel network of MOS varactors vsthe input tuning signal, with one terminal of every MOS varactor beingbiased to the same input tuning signal, the other terminal of each MOSvaractor being biased at a different level that is independent of theinput tuning signal.

In FIG. 2 a potential divider circuit is illustrated comprising an arrayof resistors R1 to R17 connected in a ladder arrangement. Resistors R1to R6 are connected in a series chain to form a first side X andresistors R12 to R17 are connected in a series chain to form a secondside Y. Resistors R7 to R11 are each connected between the respectiveresistors of the first and second chains X and Y to form the centralrungs Z of the ladder arrangement.

Control signals for variable reactors are derived from the potential ateach of nodes N1 to N7 of the first side X. The ladder may be longer orshorter depending upon requirements.

Input applied signals are applied to the feet and tops of the ladder butillustration control potentials V0, V1A, V1B, V2A, V2B, V3A, V3B areshown.

These input applied signal components are independent of the inputtuning signal:

-   Potential V0 serves to apply a fixed offset frequency;-   Potentials V1A and V1B are equal, and may be used to adjust the    sensitivity of tuning;-   Potentials V3A and V3B are equal, and may be used to adjust    polynomial components of the frequency response to the input tuning    signal up to the second order;-   Potentials V3A and V3B are equal, and may be used to adjust    polynomial components of the frequency response to the input tuning    signal up to the third order.

The “purity” of the adjustment will depend on the values of theresistors and the sizes of the variable reactors; and the third-orderadjustment will inevitably affect nominal sensitivity. As the inputtuning signal independent potential components on the control portslocally affects the inverse of the tuning sensitivity, higher ordercomponents will be increasingly present as the adjustment increases(except zero and possibly first order).

However, control voltage components will not necessarily be independentof the input tuning signal. For example, the tuning range can beincreased by making V0 dependent on the input tuning. An extension ofadjustment to the fourth order may be achieved by applying a componentthat is (linearly) dependent on the input tuning signal to V3A and V3B.

If adjustment is only required up to the second order, this may beachieved using a simple resistor chain (i.e. omitting R7, R8 . . . andV2A, V2B . . .), and applying a component that is (linearly) dependenton the input tuning signal to V1A and V1B.

Thus the circuit of FIG. 2 provides for adjustment of different ordersof the tuning characteristic. Extension to higher orders isstraightforward, for example, by generating input tuningsignal-independent components that have second and higher orderdependence on their position in the resistor chain on the right side ofthe diagram. This may be achieved, for example, by attaching furtherhalf-ladders to the right hand side of the ladder to create arectangular grid of resistors. Alternatively, fourth order adjustmentmay be implemented within the present array by including a term that islinearly dependent on the input tuning signal in the V2A and V2B voltagesources. This is just one example of a resistor arrangement that canprovide these characteristics. Similar arrangements for adjusting thebias differentials and linearity within such chains can also be usefulto modify the references for gradient changes of a piece-wise-lineargenerator.

Individual different potentials are taken from the left hand side of thediagram to form the different value individual control voltages to beapplied to variable capacitor elements such as shown in FIG. 3.

FIG. 3 is a circuit diagram showing one possible application of themethod of the present invention. The circuit is shown as used for thesimulation that generated FIG. 1 and thus includes voltage and currentsources V_(in), I₁, and I₂ and V_(diode) as used to confirm performanceduring simulations. The circuit comprises a segmented arrangement MOSvaractors Q1 to Q18 arranged in a chain of back-to-back pairs. The areasof each of the MOS varactors are indicated by the area multiplier M. Inthis embodiment the two varactors of each back-to-back pair have thesame area as each other but the varactors of different pairs havedifferent areas. Thus, in this example, each of Q1 and Q6 have an areagiven by a width W of 5 mm and a height H of 5 micrometers, i.e. 25square micrometers, whereas the adjacent pair of varactors Q18, Q17 hastwice this area (M=2) and the next but one pair each has an area of 20times this area (M=20).

In this particular example the gates of the MOS varactors are biasedwith uniform separation at the nodes of resistors R10, R20, R30, R40,R90, R110, R120 and R100, as explained below, and a common controlsignal is applied to the varactor wells from source Vc. Vc is applied tothe varactors on the left side of the pairs via a 1 megohm resistor R50and to the right side via a 1 megohm resistor R140. A voltage bias V_(B)is applied to the gates of each of the varactors. Under theseconditions, a small modulation of the position-dependent bias can causea change in the modulation function of the MOSFET varactor assembly thathas a similar (but normally inverse) characteristic.

The control voltage potentials generated by the divider of FIG. 2, thetuning signals VA, VB and VC and the sizes and characteristics of thevaractors determines the impedance which the arrangement presents to thecircuit to be tuned, such as an oscillator or filter circuit.

FIG. 4 a shows a specific example of the general resistor bridgearrangement of FIG. 2, which can be used to control theposition-dependent bias on the varactor chain of FIG. 3.

This example of a resistor arrangement comprises a first series chain Xof 1kΩ resistors R101 to R112, a second series chain Y of 1kΩ resistorsR201 to R211 and an array Z₁ of 20kΩ bridging resistors R301 to R311,connecting corresponding 1kΩ resistors in the first and second serieschains, as shown.

Each of the outputs at voltage nodes V0 to V12 will be different anddependent upon its position number. Applied voltages are shown at VA, VBand VC, as in FIG. 2. VA is the primary applied voltage applied acrossthe whole bridge array between the node corresponding to V₀ and the nodecorresponding to V₁₂. VB is applied between the node N₀ corresponding toV₀ and the node N₆. VC is applied between the node N₁ and the node N₁₁.

If the voltage VB applied to node N6 is half the primary applied voltagei.e. VB=0.5VA, and voltage source VC applied between nodes N1 and N11 isfive sixths of the primary voltage source VA, i.e. VC=5×VA/6 then thepotential on the nodes will vary linearly with position number. Ifsource VB is changed from 0.5×VA then it will generate an additionalsecond-order position-dependent term, leaving the bias at voltage nodesV₀ and V₁₂ unchanged. Similarly, a modification to source VC willgenerate a third-order position-dependent term that leaves the bias atvoltage nodes V₀, V₆, and V₁₂ unchanged. FIG. 4 b shows the effect ofmodifying sources VB and VC respectively by 2.36 and 18.7 voltsrespectively, and it will be seen that, even for the relativelyclosely-spaced resistor values of FIG. 4 a, the peak distortion of thesecond-order and third-order terms is only 0.33% and 3.5% respectively.For particularly close tolerance applications this distortion levelcould be corrected with minor modifications to the resistor values.

FIG. 5 shows an alternative resistor topology to that of FIGS. 2 and 4a. This also is suitable for reference chains with adjustablesensitivity and distortion.

This embodiment comprises a series chain of resistors R501 to R512biased by the main voltage source VA. Each voltage node V₁ to V₁₁between the respective resistors R501 to R512 in the chain is connectedto one terminal of each of respective resistors R521 to R531 on one sideand R541 to R551 on the other side. The other terminals of resistorsR521 to R531 are all connected together to the common source VE and theother terminals of resistors R541 to R551 are all connected together tothe common source VD.

In principle, the first, second and third order dependencies of thevariation of potential with position may be controlled as follows:VD=V1/2+V2+V3VE=V1/2+V2−V3Where V1 is the linear term, V2 provides even-order terms, and V3provides odd-order terms. This topology has the advantage of requiringlower dissipation to provide a given output impedance. The disadvantageis that the resistor values are not simply related, which complicatesthe layout and makes it harder to attain accurate proportional matching.Note that the values presented in FIG. 5 give good performance for thelinear term only—further adjustment may be needed to provide lowdistortion for the even-order and odd-order correction terms to providelow content beyond the third order.

In the embodiments described above, each of the individual MOS varactorsgenerally sees the same input tuning signal change. However it willsometimes be advantageous to drive some of the end sections with morerapid gradients. This may mean that the drive to these end sectionsneeds to become non-linear over some part of the range—but the effectneed not be significant if the region of reduced gain coincides withreduced tuning effect from that section. A particular application wouldbe where it is difficult to maintain adequate and uniform tuning rightup to the end of the tuning range.

Suitable gradients for use in the invention are illustrated with thegraphs of FIGS. 6 to 13 for the case of linearity. In all of FIGS. 6 to13 the horizontal axis represents the input tuning signal; the verticalaxis represents the potentials applied to the MOS varactors' controlports for FIGS. 6, 8, 10, and 12, and the control signal applied to theMOS varactors for FIGS. 7, 9, 11, and 13.

FIG. 6 shows signals which may be applied to the control ports of thetuning circuits, i.e. the MOS varactors in an embodiment in which thecontrol signals are to be independent of the input tuning signals.Signal 600 is applied to one of the ports of each of the tuning circuitsand each of signals 601 to 606 are applied to the other port of therespective tuning circuit. As can be seen, signals 601 to 606 areindependent of the input tuning signal which is in the range of 0 to5.1V. The supply range is 0 to 5.5V.

FIG. 7 illustrates the equivalent composite control signals, i.e. thedifference between the signals applied to the control ports forming thecontrol input of each tuning circuit. Line 700 represents theapproximate signal level region where the tuning sensitivity peaks,whereas line 710 represents the onset of significant tuning sensitivity,i.e. if the signal gradient reduces or becomes zero below this line, itwill make little difference to the overall sensitivity.

FIGS. 8 and 9 illustrate the signals when signals 801 to 806, areapplied to one port of the control input of each of the tuning circuits,and signal 800 is applied to the other input port of each tuningcircuit, to produce the composite respective signals 901 to 906. Lines900 and 910 are equivalent to the lines 700 and 710 in FIG. 7. Thisincreases the tuning range a modest amount.

FIGS. 10 and 11 illustrate a single-ended arrangement, with potentialsindependent of the input tuning signal, which requires limiting when thesignal on an input port would otherwise be near or below zero. One portof all the control inputs of the variable reactors is connected to acommon terminal of the oscillator, the potential of which is shown bythe dashed line. This can explore much of the available control rangeand provide an input tuning signal dependent control component to thesecond control port of the majority of the control inputs, except theone with the lowest potential. This increases the control range on mostports by about 0.9V. Lines 1100 and 1110 are equivalent to lines 700 and710 in FIG. 7. Clearly, such a scheme will provide the maximum lineartuning range that can be achieved using control signals that arelinearly dependent on the input tuning signal throughout their activeranges. However, we can see that neither the highest signal shown northe lowest explore all the useful range between line 110 and the maximumavailable potential. Thus, we could in principle extend the lineartuning range by allowing the gradients of the control signals with thehighest and lowest potentials to become steeper at the ends of thecontrol range.

FIGS. 12 and 13 illustrate a double-ended arrangement providing themaximum practical linear tuning range for control signals that arelinearly dependent on the input tuning signal throughout the operationalrange

FIG. 12 shows the control potentials applied to the control ports, andFIG. 13 shows the resultant control potentials. Again lines 1300 and1310 are equivalent respectively to lines 700 and 710 in FIG. 7.

In FIG. 12, the potential exhibiting the steepest positive gradient withrespect to the input tuning signal cannot be applied to the port thatgenerates the control signal occupying the lowest position in FIG. 13,as the tuning gradient will undoubtedly reduce once all the controlpotentials are above line 1300.

In some embodiments of the invention, the tuning circuits have twocontrol ports, such that the effective control signal is the differencebetween the signals applied to those ports. A single control signaldependent on the input signal is applied to one (same polarity) port ofall the tuning circuits. Signals that are independent of the inputsignal are applied to the other control ports of the tuning circuits.Thus, generation of the control signal from the input signal isrelatively straightforward, and distinguished only by atuning-signal-independent offset between them. This also has someperformance advantage.

A variation results in the need to adjust the characteristics to providea specific sensitivity with adequate linearity. Resistor networks thatcan provide voltage components that are orthogonal polynomial functionsof the connection position of the respective tuning circuit can bedeveloped. Each polynomial function can be modified by varying thepotential to a single terminal of the network. This allows thesensitivity and the linearity of the tuning to be conveniently adjustedby applying potentials that are independent of the input signal to thenetwork.

In embodiments using MOS varactors, a typical N-WELL varactor may beused, with the n-well characterised as the positive terminal. For biasbelow a certain level, the capacitive impedance becomes constant. As thebias rises above a threshold, the capacitive impedance starts to rise.The rate of change of impedance with voltage rapidly reaches a peak, andthen relatively gradually diminishes. For higher bias voltages, theimpedance varies approximately with the square root of bias voltage.

As far as the RF signal is concerned, groups of MOS varactors areconnected in parallel. At one extreme of the input signal, the MOSvaractors are biased with one varactor near the peak sensitivity, theothers below this point, so that some of them show virtually no changewith bias voltage initially. As the voltage rises, and the sensitivityof the first varactor reduces, first one and then the other varactorssuccessively go through their regions of maximum sensitivity. As thelast of the varactors approaches its region of maximum sensitivity, thelinear tuning range of the arrangement becomes exhausted.

One consequence of this is that the majority of the tuning circuits onlysee a fraction of the maximum available bias potential, which limits thetuning range.

The two control ports are decoupled at DC from the resonant circuit. ForIC use, the decoupling capacitors take up additional space; in addition,their series impedances can have the effect of reducing the effectivetuning range. In the present system, an on-chip supply voltagemultiplication may be provided to enhance the tuning range.

For applications that require a large tuning capacitance, and are(relatively) easy to tune, an arrangement can be used where one end ofthe varactor is connected directly to the oscillation circuit, withconsequent fixed DC component. The control port for each tuning MOSvaractor will be driven with (different) control components that haveidentical gradients, except that they will limit when the control signalis outside the sensitive range of the tuning circuit.

For some applications a smaller capacitance may be needed and for this adevice can be connected as above, but with the control potentials thatexplore the maximum possible ranges. Another device is then placed inseries “back-to-back” with this device; these will see this “maximumexcursion” potentials on one port; but control components can be appliedwith different (reverse) gradients to the other terminal of theseadditional devices, further extending the control range.

Thus embodiments of the invention can be used to avoid the reduction inavailable tuning range that can be caused by using series isolation (egcapacitors). Eliminating series capacitors potentially also reduces thedie area needed for the circuit.

Preferably the various control circuits can sequentially be brought totheir regions of peak sensitivity through the range of the input tuningsignal, and the maximum practical tuning range can be explored byensuring that the gain from the input tuning signal to each controlsignal is sufficient to explore the available tuning range. Thus, thegain required between the input tuning signal and each control signalwill depend on the level of input tuning signal at which its associatedtuning circuit reaches its maximum sensitivity.

We can achieve this by modifying the arrangement to apply controlcomponents that are dependent on the input tuning signal to the portsthat previously used control signals that were independent of the inputtuning signal. The dependence would be inverted with respect to theoriginal dependent signal that is applied to the other port. Thesesignals could be applied via the potential divider section that wasoriginally used to set up the non-dependent differentials, such that thetuning circuit that is in its sensitive range throughout the range ofthe input tuning signal sees the lowest gain, and so on.

In this way the signal on one of the control inputs of each of thetuning circuits is determined by the resonant circuit or oscillator towhich it is attached. Effectively, each tuning circuit has a singlecontrol input. Because single-ended signals are applied, the controlsignals cannot be precisely replicated without exceeding the signalranges that were required.

1. A tuning arrangement for a resonant circuit, the tuning arrangementhaving an output reactance dependent upon a plurality of input appliedsignals, one of which is an input tuning signal, the tuning arrangementcomprising: an array of tuning circuits connected in a network whoseoutput reactance is used to control a resonance frequency of theresonant circuit, each tuning circuit having a control input and havinga reactance which varies in dependence upon a value of a control signalapplied to the control input; means for generating a plurality ofcontrol signals for application to the control inputs of the tuningcircuits; and the control signal for each tuning circuit varyingsubstantially linearly with the input tuning signal throughout apredetermined range specific to that control signal so that a frequencyresponse of the resonant circuit to the input tuning signal issubstantially linear throughout a desired range of the input tuningsignal, wherein the means for generating the plurality of controlsignals comprises a resistive network in the form of a ladder, whereinthe plurality of input applied signals are applied to the top and bottomof the ladder, and the plurality of control signals are derived frompotentials on one side of the ladder, resistances in the ladder beingarranged such that proportions of the individual input applied signalsthat are derived at control ports of the ladder result in the creationof adjustments to a tuning response that are polynomial functions of aninput tuning signal level, these polynomial functions beingquasi-orthogonal functions of a third order or less for the plurality ofinput applied signals that are independent of the input tuning signal,and application of the plurality of input applied signals that aredependent on the input tuning signal to the top and bottom of the ladderresults in a modification to the tuning response that has a powerdependency that is one order higher than said third order or less,whereby the tuning response to the input tuning signal can be adjustedand set to be substantially linear.
 2. The tuning arrangement accordingto claim 1, wherein the network is a parallel network.
 3. The tuningarrangement according to claim 1, wherein at least one said controlinput has more than one port, and wherein the one of the plurality ofcontrol signals applied to the at least one said control input isdependent upon the input tuning signal.
 4. The tuning arrangementaccording to claim 1, wherein gradients of the substantially linearfrequency response of the resonant circuit to the input tuning signalare substantially reduced in a region where a tuning gradient of theresonant frequency with respect to a respective one of the plurality ofcontrol signals is small compared with a peak gradient of the resonantfrequency with respect to that respective control signal.
 5. The tuningarrangement according to claim 1, wherein gradients of the substantiallylinear frequency response of the resonant circuit to the input tuningsignal are substantially the same, each said gradient including theeffect of any dependency of the other input applied signals on the inputtuning signal.
 6. The tuning arrangement according to claim 1, whereinthe input tuning signal is developed from a prior tuning signal, theresponse of the input tuning signal being non-linear with respect to theprior tuning signal for signals outside the range where the tuningresponse is linear with respect to the input tuning signal, thenon-linear response being such as to extend a linear range of the tuningresponse with respect to the prior input signal.
 7. The tuningarrangement according to claim 1, further comprising means for adjustingthe input tuning signal using digital information stored in anonvolatile electronic memory.
 8. The tuning arrangement according toclaim 7, wherein the adjusting means maintains constancy of an overalltuning response of the resonance frequency to the input tuning signal.9. The tuning arrangement according to claim 1, further comprising meansfor adjusting the response of the control signals to any of theplurality of input applied signals using digital information stored in anon-volatile memory.
 10. The tuning arrangement according to claim 1,further comprising means for modifying at least one of the input appliedsignals as a function of temperature to correct for temperaturedependence.
 11. The tuning arrangement according to claim 1, wherein anumber of tuning circuits in the network is substantially more than theplurality of input applied signals.
 12. The tuning arrangement accordingto claim 1, wherein the array of tuning circuits comprise MOS Varactors.13. The tuning arrangement according to claim 1, wherein a gain from theinput tuning signal to at least some of the control inputs isindividually pre-adjustable.
 14. The tuning arrangement according toclaim 1, further comprising means for modifying at least one of theplurality of input applied signals such as to maintain constant tuningfrequency response of the resonant circuit to the input tuning signalover the desired tuning range and/or over a desired temperature range.15. The tuning arrangement according to claim 14, wherein the means forgenerating the plurality of control signals comprises a distributionnetwork to which the plurality of input applied signals are applied toadjust a relative distribution of gains from the plurality of inputapplied signals to the plurality of control signals.
 16. The tuningarrangement according to claim 15, wherein the resistive networkcomprises a resistor network having a plurality of control points andhaving resistor values that are defined so that signals that arelinearly dependent signals on the plurality of input applied signals maybe applied to the plurality of control points within the resistornetwork such that one of the plurality of input applied signals adjustsfrequency offset, another one of the plurality of input applied signalsadjusts at most offset and linear variation of tuning sensitivity withsignal level, and that, for each order of response to be adjusted, thereis one of said plurality of control points that has a response up tothat order, but has substantially no response for higher orders.
 17. Thetuning arrangement according to claim 16, wherein the adjustment to thelinearly dependent signals at the plurality of control points of theresistor network comprise offsets that are independent of the level ofthe input tuning signal.
 18. The tuning arrangement according to claim16, wherein the linearly dependent signals at the plurality of controlpoints include components that are linearly dependent on atemperature-dependent signal.